Transmitter including a composite amplifier

ABSTRACT

A composite amplifier includes a main power amplifier ( 10 ) and an auxiliary power amplifier ( 12 ), which are connected to a load ( 14 ) over a Doherty output network ( 16 ). A non-linear function ( 18 ) and a cross-coupling filter ( 22 ) emulate the non-linear behavior of the output current of the auxiliary power amplifier ( 12 ) and the resulting emulating signal is subtracted from the input signal to said main amplifier ( 10 ).

This application is a continuation of international applicationPCT/SEO1/01419 filed 19 Jun. 2001 which designates the U.S.

TECHNICAL FIELD

The present invention relates to a composite amplifier of the type thatincludes a main power amplifier and an auxiliary power amplifier, whichare connected to a load over a Doherty output network. The inventionalso relates to a transmitter including such an amplifier.

BACKGROUND

In cellular base stations, satellite communications and othercommunications and broadcast systems, many radio frequency (RF)carriers, spread over a large bandwidth, are amplified simultaneously inthe same high power amplifier. For the power amplifier this has theeffect that the instantaneous transmit power will vary very widely andvery rapidly. This is because the sum of many independent RF carriers(i.e. a multi-carrier signal) tends to have a large peak-to-averagepower ratio. It also tends to have a similar amplitude distribution asbandpass filtered Gaussian noise, which has a Rayleigh distribution.

The main difficulties in a PA are efficiency and linearity. Aconventional class B power amplifier exhibits maximum DC to RF powerconversion efficiency when it delivers its peak power to the load. Sincethe quasi-Rayleigh distribution of amplitudes in the summed transmitsignal implies a large difference between the average power and the peakpower, the overall efficiency when amplifying such a signal in aconventional class B amplifier is very low. For a quasi-Rayleighdistributed signal with a 10 dB peak-to-average power ratio, theefficiency of an ideal class B amplifier is only 28%, see [1].

The linearity of an RF power amplifier is usually characterized by itsAM—AM (AM=amplitude modulation) and AM-PM (PM=phase modulation)distortion characteristics. Non-linearities manifest themselves ascross-mixing of different parts of the signal, leading to leakage ofsignal energy into undesired frequency bands. By restricting the outputsignal to a smaller part of the total voltage swing of the poweramplifier, the linearity can be increased. However, this reduces theefficiency of the amplifier even further. The linearity of a poweramplifier is also greatly reduced if the amplifier saturates (the outputvoltage is clipped). This means that it is not possible to increaseefficiency by driving the amplifier into saturation, since thedistortion will then reach unacceptable levels.

One way of increasing the efficiency of an RF power amplifier is to usethe Doherty principle [1, 2, 3]. The Doherty amplifier uses in its basicform two amplifier stages, a main and an auxiliary amplifier (alsocalled carrier and peaking amplifier, respectively). The load isconnected to the auxiliary amplifier, and the main amplifier isconnected to the load through an impedance-inverter, usually a quarterwavelength transmission line or an equivalent lumped network.

At low output levels only the main amplifier is active, and theauxiliary amplifier is shut off. In this region, the main amplifier seesa higher (transformed) load impedance than the impedance at peak power,which increases its efficiency in this region. When the output levelclimbs over the so-called transition point (usually at half the maximumoutput voltage), the auxiliary amplifier becomes active, driving currentinto the load. Through the impedance-inverting action of the quarterwavelength transmission line, this decreases the effective impedance atthe output of the main amplifier, such that the main amplifier is keptat a constant (peak) voltage above the transition point. The result is asubstantially linear output to input power relationship, with asignificantly higher efficiency than a traditional amplifier.

The transition point can be shifted, so that the auxiliary amplifierkicks in at a lower or higher power level. This can be used forincreasing efficiency for a specific type of signal or a specificamplitude distribution. When the transition point is shifted, the powerdivision between the amplifiers at peak power is shifted accordingly,and the average power loss in each amplifier also changes. The lattereffect also depends on the specific amplitude distribution.

The Doherty concept has also been extended to multi-stage (more than oneauxiliary amplifier) variants [1, 4, 5]. This allows the efficiency tobe kept high over a broader range of output power levels and varyingamplitude distributions. Alternatively, the average efficiency for aspecific amplitude distribution and a specific power level can be madehigher.

The original Doherty amplifier used a quarter wavelength transmissionline coupled directly between the outputs of the two amplifiers.However, state of the art RF power transistors require a very low loadimpedance, which means that the quarter wavelength transmission line forthe original Doherty configuration also has to be designed at acorrespondingly low impedance. A solution for this problem is given in[3] and [6] and used in [7]. This solution places the impedance inverterbetween higher impedance points, obtained through single or multiplequarter wavelength impedance transformers.

The Doherty amplifiers are known to be non-linear, and to have alinearity “inversely proportional to their efficiency” [7], especiallyoutside a narrow frequency band. Attempts have been made to reduce thedistortion and increasing the useful bandwidth by paralleling multipleDoherty amplifiers with different impedance inverter center frequencies,different bias for the auxiliary amplifiers and different matchingstructures, in order to “randomize” the inter-modulation products asmuch as possible [7]. This technique also involves complicated trimmingof bias levels.

Detailed analysis shows that a Doherty amplifier, even when made fromideal components, is non-linear for all but very narrow frequency bands.The results further show that losses, that would not affect linearity ina regular class B, A or AB amplifier, cause severe non-linearity in aDoherty amplifier. Furthermore, losses can decrease efficiency more in aDoherty amplifier than a regular amplifier (although the resultantefficiency is still higher for the Doherty), since they can cause themain amplifiers to work non-optimally in addition to just adding losses.A more detailed discussion of these effects will be given below.

Another important feature is that Doherty amplifiers are inherentlyband-limited, since the impedance inverting network only provides 90degrees of phase shift at a single frequency. This band-limiting hasseveral effects.

One important effect is that the output is distorted at frequencies awayfrom the center frequency. This effect, which severely limits the use ofthe Doherty amplifier in wideband linear applications, is due to thegrowing (chiefly reactive in the lossless case) impedance of the quarterwavelength network at frequencies away from the center frequency. Thisdistortion is present even if all components are linear and lossless,since it is due to the reflection (because of the non-zero impedance) ofthe non-linear current from the auxiliary amplifier at the impedanceinverter. The resulting voltage shows up as a stronglyfrequency-dependent non-linear component in the amplified output signal.

Another effect is that the Doherty principle, i.e. the suppression of RFvoltage rise at the main amplifier above a certain transition point,works poorly outside a limited frequency band. This is because thesuppression requires the voltages from the main amplifier and theauxiliary amplifier to be in perfect anti-phase at the output of themain amplifier. Since the quarter-wave network is really only a quarterwave (90 degrees) phase shift at the center frequency, and shorter orlonger at frequencies below and above the center frequency,respectively, this requirement gets more and more violated the furtherone gets from the center frequency of the impedance inverter.

Furthermore, the output signal is bandpass filtered through reflectionsfrom the quarter-wave network.

Losses in the transistors, impedance inverters and the DC feed networksalso give rise to unexpected distortion. This is because these lossesmake the impedance at the impedance inverter, as seen from the auxiliaryamplifier, resistive instead of the ideal short-circuit (a losslessquarter wavelength transmission line loaded with the infinite impedanceof a current generator is a short-circuit at center frequency). A finiteresistance at the output of the main amplifier, as well as losses in thequarter-wave network will cause distortion. The distortion in the outputcaused by these losses are due to the same type of reflection (but nowresistive instead of reactive) of the non-linear current from theauxiliary amplifier at the impedance inverter which causes thefrequency-dependent distortion mentioned earlier.

Losses will also possibly further decrease efficiency, since the voltageat the main amplifier will not be at its maximum at output levels abovethe transition point. By providing more current from the main amplifier,this problem can be reduced. The voltage at the main amplifier will theninstead be governed by saturation, which will lead to non-linearity inthe output. By carefully adjusting the transition point and outputcurrent from the auxiliary amplifier (by adjusting the bias level andgain of the drive signal) the output can again be made more linear (atleast decreasing the amplitude distortion). This last effect is due tothe increased impedance at the output of the auxiliary amplifier, whichmakes the auxiliary amplifier contribute more voltage to the output foreach unit of current provided. The trimming method just described onlyworks in a narrow band and is not easily reproducible since it involvesusing the saturation non-linearity, whose exact shape now becomesimportant. Due to non-linear coupling to generated overtones it can alsogive a high and unpredictable AM-PM distortion.

The non-linear characteristic of the regular Doherty amplifier built andoptimized with the techniques mentioned is highly complex. It is anon-linearity whose AM—AM and AM-PM distortion varies strongly withfrequency and has a frequency (filter) characteristic that variesnon-linearly with amplitude. This makes it very difficult to compensatefor by applying pre-distortion. Since the pre-distorter would have to bevery complex (and hence implemented with digital signal processingtechniques), and a pre-distorter has to have a rather wide bandwidthcompared to the already distortion-widened signal it should compensatefor (since the inverse function to the distortion function is of higherorder than the distortion function itself, such a pre-distorter would behard to build even for moderately wideband signals.

The conclusion is that the current way of building Doherty amplifierscan only provide reasonable linear performance and efficiency in anarrow band, and this only by relying on saturation effects in the mainamplifier. Furthermore, the nonlinear characteristic is not easilycompensated for in a wide band by using pre-distortion.

SUMMARY

An object of the present invention is to enhance linearity of acomposite amplifier provided with a Doherty output network, preferablyover a broad frequency band.

This object is achieved in accordance with the attached claims.

Briefly, the present invention subtracts a non-linear function of theinput signal, which emulates the non-linear auxiliary amplifier outputcurrent, from the main amplifier drive signal. This has the advantage ofcanceling the non-linear components in the output without sacrifyingamplifier efficiency.

The non-linear function can be obtained from a model of the auxiliaryamplifier current function (if the auxiliary amplifier is producing thenon-linear current by working in class C), or can be produced beforehandand used, in amplified form, both as the drive signal for the auxiliaryamplifier (which then can be biased for linear class B or AB operation)and for cross-coupling through the filter.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention, together with further objects and advantages thereof, maybest be understood by making reference to the following descriptiontaken together with the accompanying drawings, in which:

FIG. 1 is a simplified block diagram of an exemplary embodiment of thecomposite amplifier in accordance with the present invention;

FIG. 2 is a model of the output network of a Doherty amplifier;

FIG. 3 a diagram illustrating the frequency dependence of the impedanceseen by the auxiliary amplifier output;

FIG. 4 is a diagram illustrating the frequency dependence of thetrans-impedance between the main and auxiliary amplifier;

FIG. 5 is a simplified block diagram of another exemplary embodiment ofthe composite amplifier in accordance with the present invention;

FIG. 6 is a diagram illustrating the input-output voltagecharacteristics of a prior art Doherty amplifier;

FIG. 7 is a diagram illustrating the input-output voltagecharacteristics of a composite amplifier in accordance with the presentinvention;

FIG. 8 is a block diagram of a Doherty output network with LDMOStransistors and matching to 50 ohms;

FIG. 9 is a model of the output network of a multi-stage Dohertyamplifier;

FIG. 10 is a simplified block diagram of a multi-stage embodiment of thecomposite amplifier in accordance with the present invention;

FIG. 11 is a diagram illustrating the input-output voltagecharacteristics of a multi-stage composite amplifier in accordance withthe present invention;

FIG. 12 is a block diagram of a simplified embodiment of the compositeamplifier in accordance with the present invention;

FIG. 13 is a diagram illustrating the input-output voltagecharacteristics of the simplified composite amplifier in accordance withFIG. 12;

FIG. 14 is a simplified block diagram of a another exemplary embodimentof the composite amplifier in accordance with the present invention;

FIG. 15 is a simplified block diagram of another exemplary embodiment ofthe composite amplifier in accordance with the present invention;

FIG. 16 is a simplified block diagram of a another exemplary embodimentof the composite amplifier in accordance with the present invention; and

FIG. 17 is a simplified block diagram of another exemplary embodiment ofthe composite amplifier in accordance with the present invention.

DETAILED DESCRIPTION

The basic principles of the present invention will now be described withreference to FIG. 1-4.

FIG. 1 is a simplified block diagram of an exemplary embodiment of thecomposite amplifier in accordance with the present invention. Thisembodiment illustrates the basic principles of the present invention. Itincludes a main power amplifier 10 and an auxiliary power amplifier 12.The output of auxiliary amplifier 12 is connected directly to a load(antenna) 14, whereas the output of main amplifier 10 is connected tothe output of auxiliary amplifier 12 over a Doherty output networkincluding a quarter wavelength transmission line 16. On the input sidean RF (Radio Frequency) input signal x is divided into two branches, onebranch intended for main amplifier 10 and another branch for auxiliaryamplifier 12. The auxiliary amplifier branch includes a non-linearfunction block 18, which transforms input signal x into f2(x), and aphase shifter 20, which shifts the input signal to auxiliary amplifier12 by 90 degrees. Elements 22 and 24 will be described below. Asindicated by antenna 14 the composite amplifier may be part of atransmitter, for example a transmitter in a base station in a cellularmobile radio communication system.

FIG. 2 is a model of the output network of a Doherty amplifier. In thismodel the active part of the amplifier transistor outputs are modeled aslinear controlled current generators. The finite output conductances ofthe transistors, together with possible reactances, are lumped togetheras z_(p1) and z_(p2), respectively. The impedances presented to eachcurrent generator output node are defined as:$z_{11} = { \frac{v_{1}}{i_{1}} \middle| {}_{i_{2} = 0}\quad z_{22}  =  \frac{v_{2}}{i_{2}} |_{i_{1} = 0}}$

Similarly, the transimpedances, i.e. the voltage at the inactiveamplifier output in response to an output current at the activeamplifier, are defined as:$z_{21} = { \frac{v_{2}}{i_{1}} \middle| {}_{i_{2} = 0}\quad z_{12}  =  \frac{v_{1}}{i_{2}} |_{i_{1} = 0}}$

Assuming that all components are reasonably linear, superposition can beused for analyzing this model. The composite amplifier output voltage(at the antenna) is here assumed to be the same as the output voltage atauxiliary amplifier 12, although in reality there can be a feeder cable,filters, etc. separating the actual antenna and the amplifier output.The combined effect of all these elements is included in the antenna(output) impedance, ZANT.

In an ideal Doherty amplifier, the impedance z₂₂ is zero, i.e. a currentexcitation from auxiliary amplifier 12 does not cause a voltage in theoutput. This is because a quarter wavelength transmission lineterminated with an infinitely high impedance transforms into a shortcircuit. For a practical

Doherty amplifier, however, this impedance has a non-zero resistive partand a strongly frequency-dependent reactive part. The resistive part(which is also frequency-dependent) stems from the transformed, finiteoutput conductance of main amplifier 10. The reactive part comes partlyfrom possible reactive components of z_(p1), as well as the transformedz_(p2) and ZANT. Even without these reactances, however, there is afrequency-dependent reactive part due to the quarter-wave transformer,which is only a perfect quarter wavelength at a single frequency. Acomparison of the z₂₂ magnitude (normalized to an optimal loadresistance of 1 ohm) for the (ideal) lossless case and with large losses(both in the quarter wavelength transmission line and transistor outputconductance) is given in FIG. 3. The design frequency is 1 GHz and thereactive part dominates from 800 MHz to 1.2 GHz in the lossless case.The resistive part dominates in the lossy case. The transimpedances z₂₁and z₁₂ (which are equal, due to the Reciprocity Theorem) are alsoaffected by losses in that the transmission, i.e. magnitude of thevoltage at the opposite terminal for a given current stimulus, islowered. This effect can be seen in FIG. 4.

The non-zero impedance z₂₂ will “reflect” any current i₂ from auxiliaryamplifier 12 as a voltage, and this voltage will be found in the output.If i₂ were a linear representation of the desired signal, this would notbe a problem. However, in Doherty and similar amplifiers, this currentis a very non-linear function of the desired signal (due to functionf2(x)). The non-ideal impedance z₂₂ thus makes the amplifier outputnon-linear.

The idea of the present invention is to cross-couple a copy of thisnon-linear signal (i₂ filtered by impedance z₂₂) to main amplifier 10 inanti-phase, to effectively cancel the distortion at the output. Sincethe transimpedance z₂₁ is the main linear channel from main amplifier 10to the output, the compensation to the input of main amplifier 10 willlinearly transform (slightly filtered by z₂₁) into a cancellation signalin the output. The cross-coupled compensation signal must thereforeitself be compensated for the filtering effect of transimpedance z₂₁ foreverything to cancel perfectly. Thus, a cross-coupling filter 22 in FIG.1 may be represented by:z₂₂*z₂₁ ⁻¹where “*” denotes multiplication in the frequency domain or convolutionin the time domain. The cross-coupled signal is subtracted from theinput signal to main amplifier 10 in an element 24. In a digitalimplementation element 24 is an adder, in an analog implementation itmay be realized as a hybrid.

Having described the basic principle of the invention, an embodimentwith an equalized frequency response will now be described withreference to FIG. 5-7.

Since the primary function of auxiliary amplifier 12 in a Dohertyamplifier is to keep the voltage at main amplifier 10 below saturation,the frequency dependence of all signals at the output of main amplifier10 should be as flat as possible. For the linear component (whichconstitutes all of i₁ of an uncompensated amplifier) this is achieved byfiltering at the input by a filter with the frequency characteristics ofz₁₁ ⁻¹, the inverse filter of the impedance seen at the output of mainamplifier 10.

For the non-linear component due to i₂, which is filtered through thetransimpedance z₁₂, and the non-linear part of i₁ that represents thecross-coupled distortion-canceling signal, which is filtered by z₁₁, thetotal should have a flat frequency characteristic (not just inmagnitude, but also in phase). Since the non-linear component is formedby two parts, which are differently filtered, and the requirement fordistortion-cancellation at the output dictates a certain relationshipbetween the frequency characteristics of these signals, they should bothbe additionally filtered by the inverse of a special composite filter.Assuming that the raw non-linear function f2(x) has been filtered byz₂₂*z₂₁ ⁻¹ for the cross-coupled part of i₁ and by nothing for theauxiliary amplifier 12 part (except for gain), the total non-linear partis represented by:${\underset{\underset{i_{2}\quad{part}}{︸}}{{{f2}(x)}*z_{12}} - \underset{\underset{{cross} - {{coupled}\quad{part}}}{︸}}{{{f2}(x)}*z_{22}*z_{21}^{- 1}*z_{11}}} = {{{f2}(x)}*\underset{\underset{{{composite}\quad{filter}}\quad}{︸}}{( {z_{12} - {z_{22}*z_{21}^{- 1}*z_{11}}} )}}$

Thus, the extra equalizing filtering to these signals should have afrequency response as:(z₁₂−z₂₂*z₂₁ ⁻¹*z₁₁)⁻¹

So far, nothing has been said about the magnitudes of the currents andvoltages in the system except for their relation to each other. For thelossless case and at (near) the center frequency of the quarter-waveline, the traditional Doherty equations suffice. For extracting the mostpower from the chosen transistors, at least one of the transistorsshould be operating at its maximum current I_(max). The voltages at peakpower should also be the maximum allowed voltage V_(max) (possibly witha safety margin). For a class B amplifier, the optimal load R_(opt) isV_(max)/I_(max). For an ideal Doherty amplifier the optimal loadimpedance depends on the transition point α, such thatR_(o)=R_(opt)(1−α).

For transition points α below 0.5, the current i₁ should in the ideallossless, narrowband case vary linearly with the signal amplitude and beequal to I_(max)(1−α) at the peak amplitude. Current i₂ should insteadbe zero for output voltages below the transition point, and above thetransition point vary as the (normalized) amplitude minus α divided by(1−α). This means that auxiliary amplifier 12 delivers current I_(max)at peak amplitude. For transition points above 0.5 (which is veryunlikely for optimized multi-carrier cases), i₁ would instead amount toI_(max) at peak amplitude, and i₂ would maximally be I_(max)(1−α)/α.

The procedure for the lossy, wideband case is more involved. Thelimitations for the currents and voltages are the same as for thenarrowband lossless case, but the statistical nature of the widebandsignals makes it hard to obtain analytical expressions for them. Thevoltages will then depend on the bandwidth used, the amplitudedistribution and phase relations of the individual carriers of thesignal. The lossy, narrowband case can however provide a starting point,from where adjustments can be made for the specific signals encountered.

In the lossy case the filter for obtaining the linear part of i₁, asapplied to the dimensionless input signal x, will be V_(max)/α*z₁₁ ⁻¹.The physical meaning of this filter is to generate the current i₁ suchthat the voltage at the output of the current generator of mainamplifier 10 reaches V_(max) at the normalized input amplitude α whenthe impedance seen by this current generator is z₁₁. The term z₁₁ ⁻¹,the inverse filter of the impedance z₁₁, is when observed in thefrequency domain equal to 1/z₁₁.

The filters applied to the non-linear function f2(x) also have thedimension of current. In practice this is achieved by generating theappropriate drive voltage to the transistors, which act astransconductances, so that the end result is the desired current output.The filter applied to f2(x) for obtaining i₂ is in the lossless casewithout frequency compensation simply a multiplication by j*I_(max) (90degrees phase shift). The maximum amplitude of the function f2(x) ishere assumed equal to one. The cancellation term is then f2(x) filteredby −j*I_(max)*z₂₂*z₂₁ ⁻¹. The compensation (z₁₂−z₂₂*z₂₁ ⁻¹*z₁₁)⁻¹ forachieving a frequency-independent non-linear voltage at main amplifier10 can be multiplied to these two expressions in normalized anddimensionless form.

The expression for obtaining the linear part of i₁ already compensatesfor losses. The expressions for the non-linear parts must be modified todo so. Since the relation between the two non-linear currents is alreadyestablished, this is achieved by modifying the magnitude (gain) of bothparts equally, so that the amplitude of the suppression voltage at mainamplifier 10 has the same slope as the linear part. The factor tomultiply with is V_(max)/α divided by (z₁₂−z₂₂*z₂₁⁻¹*z₁₁)*j*I_(max)/(1−a). The numerator and denominator are the voltagerise per normalized amplitude for the voltage at main amplifier 10 dueto the linear part of i₁ and the non-linear currents, respectively. Thedenominator represents the voltage rise when the current magnitudederived for the narrowband, lossless case is used. One thing to notehere is that the compensation (z₁₂−z₂₂*z₂₁ ⁻¹*z₁₁)⁻¹ for achieving afrequency-independent non-linear voltage at main amplifier 10 isautomatically included in this “new” compensation. Thus, in hindsightthe normalization is actually not necessary.

The analytical expressions for obtaining i₂ and i₁ are thus:$\begin{matrix}{i_{2} = {\frac{V_{\max}( {1 - \alpha} )}{\alpha}( {z_{12} - {z_{11}*z_{22}*z_{21}^{- 1}}} )^{- 1}*{{f2}(x)}}} \\{i_{1,\quad{nonlinear}\quad{part}} = {{- \frac{V_{\max}( {1 - \alpha} )}{\alpha}}\underset{\underset{{Equalizing}\quad{part}}{︸}}{( {z_{12} - {z_{11}*z_{22}*z_{21}^{- 1}}} )^{- 1}}*\underset{\underset{{{Distortion} - {cancelling}}\quad{part}}{︸}}{z_{22}*z_{21}^{- 1}*{{f2}(x)}}}} \\{i_{1,\quad{linear}\quad{part}} = {\frac{V_{\max}}{\alpha}z_{11}^{- 1}*x}}\end{matrix}$

As previously, if the dimensionless signals f2(x) and x are representedin the time domain, “*” represent convolution in the time domain. Ifthey are represented in the frequency domain, the symbol insteadrepresents multiplication of frequency responses, and the multiplicationwith inverse filters can be written as a division by the filter instead.The j and −j factors have vanished from the expressions, but in realitythe phases of the currents are about the same as before. What hashappened is that the imaginary units are embedded into the (z₁₂−z₂₂*z₂₁⁻¹*z₁₁)⁻¹ factors. Since z₁₂ (the largest part of the expression, atleast near the center frequency) mainly represents the transformation ofa current into a voltage over a quarter-wave line, this entails a 90°phase shift at the center frequency. The derived network model is shownin FIG. 5. Filters 22, 26 and 28 may thus be represented by:$\begin{matrix}\begin{matrix}{{Filter}\quad 28\text{:}\quad\frac{V_{\max}( {1 - \alpha} )}{\alpha}( {z_{12} - {z_{11}*z_{22}*z_{21}^{- 1}}} )^{- 1}} \\{{Filter}\quad 22\text{:}\quad\frac{V_{\max}( {1 - \alpha} )}{\alpha}( {z_{12} - {z_{11}*z_{22}*z_{21}^{- 1}}} )^{- 1}*z_{22}*z_{21}^{- 1}}\end{matrix} \\{{Filter}\quad 26\text{:}\quad\frac{V_{\max}}{\alpha}z_{11}^{- 1}}\end{matrix}$

So far only the optimization of the voltage at main amplifier 10 anddistortion cancellation in the output have been studied, and expressionsfor the optimal currents have been derived. The voltage amplitude atauxiliary amplifier 12 has been left out of the discussion. This ispartly because a fixed hardware setup has been assumed, i.e. theimpedance of the quarter wave line and the load has been assumed fixed.For a lossless system this is not a serious problem, the effect ofoptimizing for flat response and optimal amplitude at main amplifier 10is that the output signal gets a slight frequency dependence. Whenlosses are considered, however, the effect can be that the maximumvoltage at auxiliary amplifier 12 never reaches V_(max), even at maximuminput levels. This constitutes a more serious problem, since thetransistors then deliver less than the maximum power to the load (atpeak output), while still having the same supply voltage, and theefficiency will drop. The simple solution is to either reduce the supplyvoltage, or to increase the load impedance until maximum voltage isachieved at peak output (the latter solution is preferred, since thisscheme gives higher efficiency and more available output power). Thecompensation for losses can also have the effect that neither transistorreaches I_(max), which also implies an under-utilization of thetransistors. Impedances (load and quarter-wave line) may then have to bechanged in order to use the maximum possible output power from thetransistors. Equally important is to keep both transistors in the saferegion, so that the maximum currents and voltages are reached but notexceeded. Note that when changing the impedances in the circuit,redesign of the compensations according to the depicted scheme isnecessary. Also, if maximum power is not a design goal, the circuit canbe optimized differently, to meet other objectives.

The effect of the compensation in accordance with the present inventionis illustrated in FIGS. 6 and 7 with reference to a simulated examplewith a multi-carrier signal. The signal consists of nine carriers within80 MHz bandwidth centered on 1 GHz.

In this example, losses are present both as losses in the quarter wavetransmission line and as conductive losses at the outputs of thetransistors. In FIG. 6 the normalized magnitudes of the voltages at mainamplifier 10 and auxiliary amplifier 12 are plotted against the desiredmagnitude (the normalized amplitude of x) for the uncompensated case(prior art). The drive signals have been adjusted to keep both voltageswithin the linear (unsaturated) range of the transistors. The differentslopes of the output signal (voltage at auxiliary amplifier 12) belowand above the transition point indicate a static non-linearity. Thedifferent widths of these curves indicate a level-variant frequencydependence. The voltage at main amplifier 10 is not at all close to thedesired constant level above the transition point, which means that theaverage efficiency will be low (although still probably better than fora class B amplifier).

The normalized magnitudes of the voltages at main amplifier 10 andauxiliary amplifier 12 after distortion-canceling andefficiency-boosting cross-coupling in accordance with the presentinvention are illustrated in FIG. 7. Compensation of the network forlosses has been performed by changing the transmission line impedanceand the load impedance. The output voltage can be seen to be linear, andthe voltage at main amplifier 10 clearly is close to optimal forefficiency. The widened lines are for both voltages due to bandwidthrestrictions, for the linear part due to the Doherty network and for thenon-linear parts due to the simulated bandwidth of about 400 MHz.

Sometimes the optimal load impedances of the transistors are muchdifferent from the impedances available for the quarter-wave line andload. The transistors are often also packaged, which means that thecurrent-source output is only available indirectly. A Doherty amplifiercan still be made, by moving the quarter-wave line to a point one or twoquarter wavelengths from the transistor by appropriate matching networks[3, 6, 7].

An example of a modified Doherty network employing Laterally DiffusedMetal Oxide Semiconductor (LDMOS) Field Effect Transistors (FETs) isshown in FIG. 8 (for the purposes of this application, such an outputnetwork will still be considered as a Doherty output network). Itconsists of two-stage matching networks closest to the transistors andthe regular quarter-wave line outside the matching networks. Eachmatching network consists of two pi-matching sections in which thecapacitor C_(m1) is a part of both the first and second, possiblysymmetrical, sections. The usually very large output capacitance CDS ofthe LDMOS transistor implies that the matching section closest to thetransistor has a very low impedance. The second section transforms thesystem impedance, usually in the order of 50 ohms, down to this level.The matching can be made equal for both branches, if a quarter-wave linecan be made that has the appropriate impedance Z_(t). Alternativelydifferent matching networks can be used depending on which load andtransmission line impedances are available.

The modified Doherty network in FIG. 8 has three nodes worth analyzing.The design of the cross-coupling distortion-cancellation signal in thiscase starts by identifying how RF currents from the node of mainamplifier 10 and the node of auxiliary amplifier 12 transform intovoltages at the output node. This yields a relation between thecross-coupled part and the “direct” part, such that the cross-coupledpart should have an extra filter of −z_(o2)*z_(o1) ⁻¹, where:$z_{02} = { \frac{v_{0}}{i_{2}} \middle| {}_{i_{1} = 0}\quad z_{01}  =  \frac{v_{0}}{i_{1}} |_{i_{2} = 0}}$

The linear part of current i₁ is also for the modified networksdetermined from the expression x*V_(max)/α*z₁₁ ⁻¹ which gives the gainand filter characteristics of this part.

The “filter factor sum” (z₁₂−z_(o2)*z₁₁*z_(o1) ⁻¹) of the non-linearvoltages at the current generator output of main amplifier 10, and the“equal slope” criterion, give the full expressions for the direct andcross-coupled filters. The procedure is similar to the one derived forthe simple Doherty network, except that the new transimpedances z_(o2)and z_(o1) are used instead of z₂₂ and z₂₁. Something to note,especially for the modified networks, is that the narrow bandwidth cancause problems for the cancellation operation. Since(z₁₂−z_(o2)*z₁₁*z_(o1) ⁻¹) can have zeros not very far from the centerfrequency, the inverse of this filter, which is applied to thenon-linear components of the currents, will have infinitely highamplitude at these points. The compensation (and hence the bandwidth ofthe non-linear signals) must therefore be limited to a sufficientlynarrower bandwidth than these “compensation poles”. Except for theseconsiderations, the analytical expressions for obtaining i₂ and thecancellation term of i₁ for the fully compensated modified Dohertyamplifier are: $\begin{matrix}{i_{2} = {\frac{V_{\max}( {1 - \alpha} )}{\alpha}( {z_{12} - {z_{11}*z_{02}*z_{01}^{- 1}}} )^{- 1}*{{f2}(x)}}} \\{i_{1,\quad{nonlinear}\quad{part}} = {{- \frac{V_{\max}( {1 - \alpha} )}{\alpha}}\quad\underset{\underset{{Equalizing}\quad{part}}{︸}}{( {z_{12} - {z_{11}*z_{02}*z_{01}^{- 1}}} )^{- 1}}*\underset{\underset{{{Distortion} - {cancelling}}\quad{part}}{︸}}{z_{02}*z_{01}^{- 1}*{{f2}(x)}}}} \\{i_{1,\quad{linear}\quad{part}} = {\frac{V_{\max}}{\alpha}z_{11}^{- 1}*x}}\end{matrix}$

The voltage at auxiliary amplifier 12 will have a different frequencydependence for the linear and non-linear parts. It is however notnecessary to compensate for this, as long as the maximum voltages andcurrents are not exceeded, since auxiliary amplifier 12 is not theoutput node. The guidelines described in conjunction with the simpleDoherty amplifier, about maximizing the available power by reaching themaximum (safe) currents and voltages but not exceeding them, hold alsofor the modified Doherty amplifier. The recipe is the same; change loadand transmission line impedances until most voltages and currents reachtheir maximum values at some point of the desired amplitude range.

A multi-stage Doherty amplifier presents yet another challenge, sinceeven more nodes are present in the system. The distortion should beminimal at the output and efficiency boosting should ideally optimizethe voltage levels above transition points for several (all except thelast one) amplifiers. The basic rules developed earlier still apply, buttrade-offs may be necessary to get the best overall result. Amulti-stage composite amplifier operating in accordance with theprinciples of the present invention will now be described with referenceto FIG. 9-11.

The main characteristic of multi-stage Doherty amplifiers is that theyhave more than one amplifier (current generator) coupled with hightransmission (transimpedance) to the output. This means that for linearoperation (i.e. without saturation or limiting effects), the linearoutput voltage even in an ideal multistage Doherty amplifier is composedof two or more non-linear parts coming from different amplifiers.

Two special arrangements are necessary for achieving linear output.Firstly, the non-linear signals that are used for depressing the voltageat the previous (one lower transition point) amplifier must at theoutput be cancelled by a similarly filtered non-linear signal from anamplifier with a high transimpedance to the output, usually the previousamplifier itself. Secondly, the non-linear parts that together make upthe linear output must have equal frequency dependence and gain, as seenat the output.

The current generators that have high transmission to the output alsohave high transmission to each other. This effect is as important forkeeping the lower power amplifiers at constant voltage above transitionpoints as the proper “Doherty effect”.

The general case of multistage Doherty amplifiers will now beexemplified with a three-stage amplifier, which is illustrated in FIGS.9 and 10. In the output network of FIG. 9 the following definitions willbe used: $z_{11} = { \frac{v_{1}}{i_{1}} \middle| {\begin{matrix}{i_{2} = 0} \\{i_{3} = 0}\end{matrix}\quad z_{22}}  = { \frac{v_{2}}{i_{2}} \middle| {\begin{matrix}{i_{1} = 0} \\{i_{3} = 0}\end{matrix}\quad z_{33}}  =  \frac{v_{3}}{i_{3}} \middle| \begin{matrix}{i_{1} = 0} \\{i_{2} = 0}\end{matrix} }}$$z_{12} = {z_{21} = { \frac{v_{1}}{i_{2}} \middle| {\begin{matrix}{i_{1} = 0} \\{i_{2} = 0}\end{matrix}\quad z_{13}}  = {z_{31} = { \frac{v_{1}}{i_{3}} \middle| {\begin{matrix}{i_{1} = 0} \\{i_{2} = 0}\end{matrix}\quad z_{23}}  = {z_{32} =  \frac{v_{2}}{i_{3}} \middle| \begin{matrix}{i_{1} = 0} \\{i_{2} = 0}\end{matrix} }}}}}$

In FIG. 10 the three non-linear functions f1(x), f2(x) and f3(x) of theinput signal are all assumed to have the same amplitude slope as thenormalized input signal x (which a slope equal to 1). The phases ofthese signals are also identical to that of input signal x.

The first function, f1(x) is equal to x below the second transitionpoint α₂. Above this point it has the same phase as x and the amplitudeis equal to a constant α₂.

The second non-linear function f2(x) is zero until the amplitude of x isat α₁, and its amplitude rises linearly from there.

The third function f3(x) behaves like the second, but starts risingabove α₂.

The first and third functions added together return the input signal x.The last statement (and generally that the sum of every other functionreturns the linear input signal) is the main requirement, even if thenon-linear functions are non-abrupt and/or shaped by polynomials of theinput amplitude or power. The shapes of the signals that are primarilydesigned to suppress voltages above transition points are only importantto the extent that they should suppress voltages good enough.

The voltage at main amplifier 10 should be as constant as possible atall levels above the first transition point α₁. As before this is partlyachieved by inverse filtering for the impedance seen at this poweramplifier by applying V_(max)/α₁*z₁₁ ⁻¹ to f1(x) for the main amplifiercurrent. For achieving a linear output, the current from secondauxiliary amplifier 12 b must transform into a signal with the samefrequency dependence at the output node as the transformed current frommain amplifier 10. This is achieved through the application of filterV_(max)/α₁*z₁₁ ⁻¹*z₃₁*z₃₃ ⁻¹ to f3(x) for the current of auxiliaryamplifier 12 b, and gives rise to different frequency dependencies forthe voltages due to currents i₁ and i₃ at the main amplifier 10 node.Since this causes a non-linear frequency dependence at this node, acompensation must be devised or otherwise the increased peak to averagenode voltage ratio would harm efficiency. A compensation can be found bytaking the difference between the frequency dependence at main amplifier10 of i₃*z₁₃ optimized for correct output and the frequency dependenceof i₃*z₁₃ optimized for flatness at main amplifier 10. This differencein filtering at main amplifier 10 is V_(max)/(α₁*(1−z₁₃*z₃₁*z₁₁ ⁻¹*z₃₃⁻¹), so the function f3(x) is filtered with this function and insertedas a part of i₂, which has high transmission to the main amplifier 10node. Since this non-linear signal would show up in the output throughthe non-ideal (ideally zero) transimpedance z₃₂, it must also be appliedto another current, preferably i₁, and both of these parts must togethercancel at the output node of auxiliary amplifier 12 b and have a flatfrequency response at main amplifier 10. Appending (i.e. multiplying)the extra filter z₃₂*z₁₁ ⁻¹ to the i₁ part and the filter(z₂₁−z₁₁*z₃₂*z₃₁ ⁻¹)⁻¹ to both parts fixes this.

The non-linear function f2(x), which ideally is applied only to i₂ andonly suppresses the voltage rise at main amplifier 10, will in thepractical uncompensated case be seen at the output due to thetransimpedance z₃₂. The compensation for this is the same as for the“difference term” compensation just described. The filter−V_(max)/α₁*(z₂₁−z₁₁*z₃₂*z₃₁ ⁻¹) ⁻¹ is applied to f2(x) for the i₂ partand the same filter without the minus sign but with an appendedfiltering of z₃₂*z₃₁ ⁻¹ is applied to f2(x) for the i₁ part.

The schematic appearance of the derived network, with the filtersdesignated by i_(ab), where a,b=1, 2, 3, is illustrated in FIG. 10. Theb's designate the function number and the a's the target poweramplifier. Since the output of the filters (in this model) are currentsand the signals fn(x), where n=1, 2, 3, are dimensionless, while thefilters have the dimension of current.

What has just been described is an optimization of the voltage at mainamplifier 10 only. This makes sense since it should have a constantamplitude for a larger part of the dynamic range than auxiliaryamplifier 12 a. The voltage at auxiliary amplifier 12 a has been leftunattended, even though it should ideally have a flat voltage amplitudefor input signals above the second transition point α₂. If losses arepresent in the circuit, load and quarter-wave line impedances can bechanged, and transition points can be moved to maximize the efficiency.If the voltage at auxiliary amplifier 12 a has not been compensated,losses can make its ideally constant level above the transition pointdroop, even if the “knee” can be made to reach the target V_(max). Thisis shown in FIG. 11.

A compensation for auxiliary amplifier 12 a above the second transitionpoint can be found by taking the sum of the voltage responses for thenon-linear functions that are sloping above this point, i.e. f2(x) andf3(x), at auxiliary amplifier 12 a. A function with this amplitude andfrequency response is then fed in anti-phase to the auxiliary amplifier12 a output node, through applying the function f3(x) through differentfilters to the currents i₁ and i₃. The current i₁ is the main channel toauxiliary amplifier 12 a, and the i₃ part is included to cancel thenon-linearity in the output. After the conditions for cancellation havebeen established (as before), the obtained filter quotient is appendedto the i₁ part and the composite frequency dependence at the auxiliaryamplifier 12 a node is calculated. The inverse of this filter term isthen appended to both parts. The result of these operations is a flatregion above the second transition point for auxiliary amplifier 12 a,at the expense of the flatness at main amplifier 10. For low-losscircuits, the main amplifier 10 node voltage is not deteriorated much,but when losses are present, it gets an upward slope above the secondtransition point. This effect probably reduces the efficiency more thanwhat is gained by having an optimal auxiliary amplifier 12 a voltage.

In the previous examples, the starting point has been to obtainfrequency-independent linear and non-linear voltages at main amplifier10. This is good for optimizing efficiency, since the flat part of thevoltage range can be held as close to the maximum as possible withoutsaturation. There are of course also other ways to obtain close tooptimum operation, which may suit a certain type of implementationbetter. Some of these will be discussed in the following paragraphs.

The principles of operation nave been described in terms of adimensionless, normalized input signal and an “end” product in the formof specially designed current outputs from the power amplifiers. Thetransistors and all other components of a practical amplifier systemhave thus been embedded into the filter equations. In reality the inputsignal can be in a variety of forms, and multiple stages of processingcan in some implementations separate the generation of the non-linearsignals, and application of the cross-coupled filters from the actualpower amplifiers. For example, the input signal can be in purely digitalform and at low frequency, if digital signal processing is used forshaping the drive functions. The transformation into voltages fordriving the power transistors (which transform their input voltage intocontrolled output currents) is then performed by a processing chaincomprising digital to analog conversion, mixers, filters and amplifiers,until the drive signals to the power amplifiers are at the rightfrequency and in the right form and size. The non-linear functions andthe cross-coupled filters will in this case be implemented entirely inthe digital domain, and may include a compensation for the frequencydependence of the up-conversion chain and transistor input matchingcircuits.

In other variants, the non-linear processing is done by non-linearcircuits at the final frequency or at an intermediate frequency. Avariety of ways to do this are available, including biasing low-powertransistors for class C operation, multiplication with a “shapingfunction” derived from the RF signal and multiplication of the linearsignal with a shaping function produced at baseband. The cross-coupledfilters can then be implemented by lumped and/or distributed filtertechniques, having current in/voltage out or voltage in/current out,being doubly terminated or any other suitable filter technique that cangive the right filter response over the desired band. The same holds forthe filters that are not cross-coupled.

In either variant, there are some basic rules that apply. Firstly, allbranches must have matched delays, i.e. the phase and time relationsbetween the different signals must be strictly controlled. Since filtersand non-linear processing have delays, any branch without a function(non-linear or filter) must be compensated by an equal delay. Thedeliberate delays that are employed to establish desired phase relationsbetween signals (i.e. quarter wave line) need not be compensated for.Secondly, the amplitude of all signals must be matched for thedistortion-cancellation and efficiency-boost to work optimally.

Although the filters may seem complicated, since they are assembled frommany frequency-dependent impedances and transimpedances, the complexityof an implementation can be reduced in several ways. In a digitalimplementation, the filters can be assembled from measured impedances bymultiplication and division in the frequency domain. The therebyassembled filters can then either be used directly for filtering in thefrequency domain, or be converted to time-domain filters. Afrequency-domain window can be applied for restricting the filters tosuitable bandwidths. Typically filters are implemented as FIR (FiniteImpulse Response) filters having a length of 20-40 taps.

As discussed in the summary section, if the auxiliary amplifier isproducing the non-linear current by working in class C, its non-linearoutput current can be modeled separately for use in the cross-coupling.The auxiliary amplifier current function can in this case not befiltered arbitrarily, since the non-linearity is in the end of theprocessing chain (in the power transistor itself). In such cases, alldistortion-cancellation is in the cross-coupled path, both the filtersand the model of the class C amplifier non-linear function. The linearpath (to main amplifier 10) can of course also have compensationfiltering in this case, as can the path to auxiliary amplifier 12,specifically to compensate for the other frequency dependencies in thispath.

The impedance of the antenna network as seen at the power amplifieroutput is generally not known in detail when producing an amplifier.However, it has an impact on the impedances in the Doherty outputcircuit. Some methods that can be used to get a better known impedanceinclude using an isolator in the antenna path, to get a more widebandresistive characteristic, or to insert a resonator or filter that ismore narrowband than the antenna network, so that the impedance of thispart (which is assumed to be reasonably known already in the productionstage) dominates instead of the actual antenna impedance.

Sometimes it is impractical to implement all of the filters that areneeded for optimal operation. There are also great differences betweenthe filters regarding how much they contribute to the overallperformance (distortion reduction and efficiency). Therefore, it can beuseful to design reduced variants with dropped or simplified filters.Generally, these reduced variants can be successful if some filter orpart of a filter can be regarded as approximately constant over thefrequency range of interest. The gain and phase value of the filter atthe center frequency can then be substituted for the fullfrequency-dependent filter. For cancellation of distortion in the outputto work, there is generally a requirement for a specific filter quotient(as previously described) between two branches. This means that aninverse of a filter, which can be hard to implement, can be dropped andthe filter itself be inserted in the other branch. Both branches mustthen be compensated for the gain and phase of the changed filter (at thecenter frequency). A very reduced variant can be found by dropping thefrequency-dependence altogether. This can possibly be useful if thefrequency range of operation is quite narrow or if losses dominate thegeneration of distortion in the output.

A simple but elegant method for obtaining the filters z₂₁ and z₂₂ is touse input-side copies of the Doherty output network, containing the samepassive circuit elements that are present in the actual output network.When such a network is driven by a current generator (small-signaltransistor) on the input side, the output voltage automatically has theright frequency dependence. The requirement for this to work is that thetransistor output parasitic elements, the quarter-wave line and theantenna network impedance can be accurately modeled. A possibility is toscale the impedance of all elements in the network to get morerealizable values and/or better voltage and current levels.

The filtering by z₂₁ can be obtained by using z₁₂ instead. In this way,the filtering of the non-linear signal by both impedances can be donewith only one copy of the output network. The drawback is that the load(over which to obtain the voltage) in this case is the model of theparasitic elements of the transistor. The load when using z₂₁ (and z₂₂)is a model of the antenna network impedance in parallel with the modelof the output parasitics of the auxiliary amplifier 12 transistor. Theantenna impedance is better known, does not spread much betweenamplifiers, and has a more convenient magnitude than the parasitics.Small-signal amplifiers with a suitable input impedance are thus easy tofind that can be complemented with reactances to form a model of theantenna network impedance.

For an implementation using only RF/microwave techniques, the simpleDoherty amplifier can be sub-optimally implemented by using the ideasfrom the previous paragraphs. The sub-optimality comes from rearrangingthe equations to make the inverse filters unnecessary, and will be shownnot to degrade the efficiency significantly from the previously derivedoptimal operation. The distortion cancellation at the output is stillcomplete. The non-linear function can (and is in this example assumedto) be generated by a class C amplifier with its bias adjusted for acertain transition point.

The inverse filter to z₂₁ is taken away from the cross-coupled path, andthis filter itself is instead inserted into the path to auxiliaryamplifier 12. The gain and phase of the filter is replaced by its valueat center frequency. The inverse of the composite filter z₁₂−z₂₂*z₂₁⁻¹*z₁₁, in the direct and cross-coupled paths is also replaced by itsgain and phase values at the center frequency, as is the filtering withthe inverse of z₁₁ in the linear path to main amplifier 10. What is leftis only the basic filtering required for perfect distortion cancellationat the output plus compensating gains for maximizing the efficiencyunder these (sub-optimal) conditions,

The simplified schematic of such a circuit is shown in FIG. 12. If thenon-linear function of the RF signal, f2(x), is produced by a class Camplifier, it can also be produced by driving amplifiers G1 and G2 inclass C mode. The signal levels in the cancellation networks are meantto be low, to minimize power consumption. The amplification to highervoltage is preferably done in the preamplifiers to main amplifier 10 andauxiliary amplifier 12.

The antenna network impedance is in this case modeled by a 50 Ohmresistance with a parallel resonator tuned to the center frequency.Amplifiers G1 and G2 are (identical) controlled current generators. Theinput impedance of (identical) amplifiers G3 and G4 together withappropriate additional reactances emulate the antenna network impedanceZANT, and possible parasitics on the output of G1 and G2 are included inthe corresponding Z_(p2) and Z_(p1). Amplifier G0 provides a matchinggain, delay and phase for the linear part to main amplifier 10. Thepower amplifiers include necessary input matching networks andpreamplifiers.

When all gains are optimally adjusted, and the output network,parasitics and antenna network impedance are properly emulated by theirinput side models, the performance will be close to optimal. The voltagemagnitudes at main amplifier 10 and auxiliary amplifier 12 for thedescribed situation are illustrated in FIG. 13. These curves have beenobtained under the same conditions and with the same bandwidths as thefully optimized case shown in FIG. 7. The voltage overshoot andundershoot at main amplifier 10 (the width of the flat part of thecurve) is only slightly higher for the non-optimal case, which meansthat the efficiency is very little degraded. An increased frequencydependence of the output signal (voltage at auxiliary amplifier 12) isseen as a widening of the voltage trace, although the amplitude part ofthis is hardly noticeable in a spectral plot.

In practice, the performance of the described methods will depend on howwell the characteristics of the Doherty output network are known.Measuring transimpedances in the output network is often hard to dodirectly, since the (RF) voltage probe and the current injector willalways have parasitics that must be taken into account. Indirectly,impedance parameters (Z-parameters) can be extracted by traveling wavemeasurements (S-parameters). A combination of different parameters thatare easy to measure can also be selected. The required filters oremulating networks can then be designed using extracted impedances andtransimpedances.

The exact values of transimpedances and impedances are hard to obtain,and are in many cases not important in themselves. Having the correctquotient of two frequency responses and the correct gains and phases inthe two paths to the common output are the most important goals forachieving distortion cancellation in the output. In the same way is thegain and phase matching of the linear part to the combined non-linearparts the most important criterion when optimization for flatness abovethe transition point. Assuming that the load and quarter-wave lineimpedances are correct, the amplitudes of flat-voltage regions and theoutput voltages can then be adjusted to maximum values by a common gainadjustment.

Using the philosophy of the previous section, a trimming scheme can bedevised. By injecting a signal at various points in the circuit, andtrimming until cancellation or another measurable condition occurs, thecircuit can be optimized for linearity and efficiency. The cancellationof distortion in the output can be handled first, by substituting awideband test signal for f2(x). For doing this under realistic operatingconditions, main amplifier 10 can be simultaneously excited by anothersignal (that is easily distinguishable in the output signal). Thefilters, gains and phases of the two canceling paths to the output canthen be trimmed, since they are fed with the same signal.

The gain of the linear path to main amplifier 10 can be adjusted (atseveral frequencies to ensure amplitude flatness) by observing thestarting point of compression in the output for a main. Compressionshould occur at a power corresponding to the transition point, if f2(x)is deactivated.

The last criterion, optimal suppression of the voltage rise at mainamplifier above the transition point, requires phase and gain matchingof the linear part to the composite non-linear part at this node. A wayto do this is to observe spectral regrowth, possibly with a simpletwo-tone test, and adjust until this is minimized. Other ways can be toadjust the phase first by cancellation in two paths to the output andadjust the gains later, knowing that the phase is correct. Either way,the phase matching, or electrical path length difference, should besufficiently correct (within a fraction of a wavelength) beforeadjustment in order to avoid local minima at multiple wavelengths awayfrom the correct one. A method of trimming while observing the spectralregrowth using a realistic (wideband, multi-carrier) signal whilesimultaneously keeping track of the power efficiency of the amplifiercan also be used.

Probing the voltage at main amplifier 10 for flatness above thetransition point, instead of just observing the efficiency, can alsohelp in achieving maximum efficiency. The probe must have high impedanceto avoid increasing the losses or otherwise detrimentally affect theconditions in the circuit. Apart from that, the probe impedance can beincorporated in the distortion-canceling and efficiency-boostingcompensations.

Many parameters of the output network and amplifiers are slowlychanging, due to aging, temperature variations and other environmentalchanges. This means that the distortion cancellation and efficiency ofthe amplifier may degrade from its initial level. To cope with thisproblem, the filters and gains in the cancellation network and thelinear path can be made to respond in real time to the parametervariations.

The adjustments described in the previous paragraphs can be automated bymonitoring the output and possibly the voltage at main amplifier 10 andrelate this to the signals inputted at various points in the network.The measured values can then be used for changing the parameters of theinput network. An alternative is to insert special signals that are onlyused for measurements (pilot signals). A pilot signal used for adjustingthe cancellation will itself cancel in the output when the input networkis properly adjusted.

An entirely digital implementation of the distortion-cancellation andefficiency-boosting techniques will have advantages over an analogimplementation in that the filters will be more accessible to detailedadjustments. An analog implementation relies on the adjustments ofcircuit elements, but the circuit itself is hard to change duringoperation.

Throughout this text, the assumption has been that saturation issomething to avoid and that the Doherty methods should be used withextra tricks to ensure that the transistors are kept out of saturation,while maximizing efficiency and minimize distortion. However, the mainreason for this is that the saturation of the lower power amplifiers,especially main amplifier 10, will otherwise occur right in the powerlevels where a multi-carrier signal statistically spends most of itstime. The distortion in the output due to a non-linearity in this regionis therefore very large. Standard power amplifiers are usually driveninto saturation in the top of their output power range, to get someextra output power and increased average efficiency. The distortioncoming from this is quite low, for an amplifier driven by amulti-carrier signal, since the probability density is exponentiallydecreasing towards the high end of the (quasi-) Rayleigh distribution ofamplitude levels. Similar methods can be used for the improved Dohertyamplifiers, by modifying the non-linear current to give a similarvoltage rise on all amplifiers above compression. Partly, this will takecare of itself when auxiliary amplifier 12 goes into saturation, so adeliberate modification might not be necessary. The delivered currentwill then be reduced, with a voltage rise also at main amplifier 10 as aresult. Some extra power, better average efficiency, and a slightlyworse distortion will be the result. The transition point can of coursealso be changed accordingly to take full advantage of this solution.Below this compression region everything will still be linear andnon-saturated.

Since saturation is a somewhat vaguely defined state, with a transitionregion in which the power amplifier is neither a pure current source nora hard-limited voltage source, solutions can be found in which a poweramplifier is held slightly saturated over the “flat” voltage range. Themethods proposed by the present invention can be used to control thisamount of saturation very precisely so that efficiency is increased,over that of a strictly non-saturated amplifier, but the distortion doesnot grow above a set limit.

The proposed solution does not preclude the use of additionallinearization techniques. After the efficiency has been optimized andthe difficult frequency dependent distortion and other large-scaleDoherty-specific distortion products has been reduced by thecancellation method, the residual distortion coming from device-specificnon-linearities can be addressed. Two popular methods for distortionreduction in wideband RF and microwave amplifiers are the feedforwardmethod and the pre-distortion method. As indicated in the backgroundsection, the reduced frequency-dependent and large-scale distortion willease the requirements for the additional linearization techniques.

An alternative to linearization loops around the whole amplifier is tolinearize the output current for each transistor (power amplifier). Thishas the added benefit that the distortion cancellation performance ofthe methods described in above can be more complete, since otherwise thenon-linearities of the current generators (transistors) will adddistortion products to the voltages that ideally should cancel in theoutput. Since these distortions are generally not equal in shape,although they can be similar in magnitude, the residual part that cannot be cancelled completely will put a limit on the cancellation of theDoherty-specific distortion.

Traditionally, Doherty amplifiers have been known to have a linearity“inversely proportional to their efficiency” [7]. The methods presentedin this document remove this tradeoff, since they can simultaneouslyoptimize the linearity and efficiency of Doherty amplifiers. Further,they can do this over very large bandwidths with retained performance.The solution is effective for all types of Doherty amplifiers and formany types of non-idealities, both those that depend on narrowbandapproximations and those due to linear parasitics.

The possibility of wider relative bandwidths and higher efficiencyenables the use of (the modified) Doherty amplifiers in previouslyunattainable areas. For example, the wider relative bandwidths makes itpossible to use the Doherty technique for radio systems at lowerfrequency, or to make high-efficiency amplifiers for entire systembandwidths instead of smaller chunks or individual channels. Even if asmaller range of bandwidth is actually used, the method enables themaking of a unified amplifier with flexible placement of the usedbandwidth or channel within a much larger bandwidth. This implies alower manufacturing cost, since fewer variants have to be manufactured.

Many different implementations are possible. Digital or analog signalprocessing can be used, and the processing can be performed with avariety of techniques, at baseband, intermediate or final (RF)frequencies. Arbitrary combinations of these can be used, matching therequirements for a function with a convenient way of implementing it.The solution can be used statically, optimized at the time ofmanufacture or at specific times during maintenance, or dynamicallyadaptive, for continuously optimizing the linearity and efficiency ofthe amplifier.

In the above description it has been assumed that a cross-coupled signalemulating the non-linear behavior of the output current of auxiliaryamplifier is subtracted from the input signal to main amplifier 10.However, as will be shown below with reference to FIG. 14-17, thecross-coupling is actually not strictly necessary. The same effect maybe accomplished in other ways.

FIG. 14 is a simplified block diagram of a another exemplary embodimentof the composite amplifier in accordance with the present invention.This embodiment is equivalent to the embodiment of FIG. 5. Thedifference is that the non-linear function 18 has been duplicated in theupper input branch to main amplifier 10. The filters are the same as inFIG. 5.

FIG. 15 is a simplified block diagram of another exemplary embodiment ofthe composite amplifier in accordance with the present invention. Thisembodiment is a simplified version of the embodiment of FIG. 14. In thisembodiment filtering is only performed in the upper input branch to mainamplifier 10. Furthermore, the filters are proportional to:

-   -   Filter 22:        z₂₂*z₂₁ ⁻¹    -   Filter 26:        z₁₁ ⁻¹

FIG. 16 is a simplified block diagram of a another exemplary embodimentof the composite amplifier in accordance with the present invention. Inthis embodiment there are different non-linear functions in the lowerand upper branches. The upper non-linear function f1(x) in block 38 hastwo constant slopes, a first slope equal to 1 up to the transitionpoint, and a second, reduced slope counteracting the non-linearity inthe lower branch. A distortion canceling filter 40 is provided only inthe lower branch. This filter is proportional to:z₂₁*z₂₂ ⁻¹

FIG. 17 is a simplified block diagram of another exemplary embodiment ofthe composite amplifier in accordance with the present invention. Thisembodiment, which is a more elaborate version of the embodiment in FIG.16, has a filter 42 in the upper branch and a filter 44 in the lowerbranch. The filters are proportional to:

-   -   Filter 44:        z₂₁*z₂₁ ⁻¹*z₁₁ ⁻¹    -   Filter 42:        z₁₁ ⁻¹

It will be understood by those skilled in the art that variousmodifications and changes may be made to the present invention withoutdeparture from the scope thereof, which is defined by the appendedclaims.

REFERENCES

-   [1] F. H. Raab, “Efficiency of Doherty RF Power Amplifier Systems”,    IEEE Trans. Broadcasting, vol. BC-33, no. 3, pp. 77-83, September    1987.-   [2] U.S. Pat. No. 5,420,541 (D. M. Upton et al.).-   [3] U.S. Pat. No. 5,568,086 (J. J. Schuss et al.).-   [4] U.S. Pat. No. 5,786,727 (B. E. Sigmon).-   [5] U.S. Pat. No. 5,025,225 (Tajima et al.).-   [6] D. M. Upton et al. “A New Circuit Topology to Realize High    Efficiency, High Linearity, and High Power Microwave Amplifiers”,    IEEE Proc. RAWCON'98.-   [7] WO 97/20385 (J. F. Long).

1. A composite amplifier including: a main power amplifier and anauxiliary power amplifier, which are connected to a load over a Dohertyoutput network; and means for emulating and compensating for thenon-linear behavior of the output current of said auxiliary poweramplifier in the input signal to said main amplifier.
 2. The compositeamplifier of claim 1, including means for equalizing the frequencyresponse of said composite amplifier.
 3. The composite amplifier ofclaim 1, including means for cross-coupling and subtracting a filteredversion of the auxiliary amplifier input signal from the main amplifierinput signal.
 4. The composite amplifier of claim 3, including across-coupling filter emulating the impedance of said auxiliaryamplifier and compensating for the transimpedance between said main andauxiliary amplifiers.
 5. The composite amplifier of claim 3, including across-coupling filter emulating the transimpedance from said auxiliaryamplifier to the output node and compensating for the transimpedancefrom said main amplifier to the output node.
 6. The composite amplifierof claim 4, including input side filters for equalizing the frequencyresponses of said main and auxiliary amplifiers.
 7. A transmitterincluding: a composite amplifier with a main power amplifier and anauxiliary power amplifier, which are connected to a load over a Dohertyoutput network; and means for emulating and compensating for thenon-linear behavior of the output current of said auxiliary poweramplifier in the input signal to said main amplifier.
 8. The transmitterof claim 7, including means for equalizing the frequency response ofsaid composite amplifier.
 9. The transmitter of claim 7, including meansfor cross-coupling and subtracting a filtered version of the auxiliaryamplifier input signal from the main amplifier input signal.
 10. Thetransmitter of claim 9, including a cross-coupling filter emulating theimpedance of said auxiliary amplifier and compensating for thetransimpedance between said main and auxiliary amplifiers.
 11. Thetransmitter of claim 9, including a cross-coupling filter emulating thetransimpedance from said auxiliary amplifier to the output node andcompensating for the transimpedance from said main amplifier to theoutput node.
 12. The transmitter of claim 10, including input sidefilters for equalizing the frequency responses of said main andauxiliary amplifiers.
 13. The composite amplifier of claim 1, whereinthe means for emulating and compensating for the non-linear behavior ofthe output current of said auxiliary power amplifier in the input signalto said main amplifier comprises: a subtractor having an output terminalconnected to an input terminal of the main amplifier; a filter having anoutput terminal connected to a first input of the subtractor, the firstfilter being an equalizing filter; a second filter having an outputterminal connected to a second input of the subtractor whereby an outputof the second filter is subtracted from an output of the first filter,the second filter being a distortion canceling and equalizing filter; afirst non-linear function having an output terminal connected to aninput terminal of the second filter; the first non-linear function andthe second filter being connected to receive a same RF signal as inputs.14. The composite amplifier of claim 13, further comprising a thirdfilter having an output terminal connected to an input terminal of theauxiliary power amplifier, the third filter being an equalizing filter;a second non-linear function having an output terminal connected to aninput terminal of the third filter; wherein the same RF signal isapplied to each of the first filter, the first non-linear function, andthe second non-linear function.
 15. The composite amplifier of claim 1,wherein the means for emulating and compensating for the non-linearbehavior of the output current of said auxiliary power amplifier in theinput signal to said main amplifier comprises: a subtractor having anoutput terminal connected to an input terminal of the main amplifier; afirst filter having an output terminal connected to a first input of thesubtractor; a second filter having an output terminal connected to asecond input of the subtractor whereby an output of the second filter issubtracted from an output of the first filter; a non-linear functionhaving an output terminal connected to an input terminal of the secondfilter; the non-linear function and the second filter being connected toreceive a same RF signal as inputs.
 16. The composite amplifier of claim15, further comprising a phase shifter having an output terminalconnected to an input terminal of the auxiliary power amplifier; asecond non-linear function having an output terminal connected to aninput terminal of the phase shifter; wherein the same RF signal isapplied to each of the first filter, the first non-linear function, andthe second non-linear function.
 17. The composite amplifier of claim 1,wherein the means for emulating and compensating for the non-linearbehavior of the output current of said auxiliary power amplifier in theinput signal to said main amplifier comprises a first non-linearfunction having an output terminal connected to an input terminal of themain amplifier, and further comprising: a distortion canceling filterhaving an output terminal connected to an input terminal of theauxiliary power amplifier; a second non-linear function having an outputterminal connected to an input terminal of the distortion cancelingfilter; the first non-linear function and the second non-linear functionbeing connected to receive a same RF signal as inputs.
 18. The compositeamplifier of claim 1, wherein the means for emulating and compensatingfor the non-linear behavior of the output current of said auxiliarypower amplifier in the input signal to said main amplifier comprise: afirst filter having an output terminal connected to an input terminal ofthe main amplifier; a first non-linear function having an outputterminal connected to an input terminal of the first filter; a secondfilter having an output terminal connected to an input terminal of theauxiliary power amplifier; a second non-linear function having an outputterminal connected to an input terminal of the second filter; the firstnon-linear function and the second non-linear function being connectedto receive a same RF signal as inputs.
 19. The transmitter of claim 7,wherein the means for emulating and compensating for the non-linearbehavior of the output current of said auxiliary power amplifier in theinput signal to said main amplifier comprises: a subtractor having anoutput terminal connected to an input terminal of the main amplifier; afilter having an output terminal connected to a first input of thesubtractor, the first filter being an equalizing filter; a second filterhaving an output terminal connected to a second input of the subtractorwhereby an output of the second filter is subtracted from an output ofthe first filter, the second filter being a distortion canceling andequalizing filter; a first non-linear function having an output terminalconnected to an input terminal of the second filter; the firstnon-linear function and the second filter being connected to receive asame RF signal as inputs.
 20. The transmitter of claim 19, furthercomprising a third filter having an output terminal connected to aninput terminal of the auxiliary power amplifier, the third filter beingan equalizing filter; a second non-linear function having an outputterminal connected to an input terminal of the third filter; wherein thesame RF signal is applied to each of the first filter, the firstnon-linear function, and the second non-linear function.
 21. Thetransmitter of claim 7, wherein the means for emulating and compensatingfor the non-linear behavior of the output current of said auxiliarypower amplifier in the input signal to said main amplifier comprises; asubtractor having an output terminal connected to an input terminal ofthe main amplifier; a first filter having an output terminal connectedto a first input of the subtractor; a second filter having an outputterminal connected to a second input of the subtractor whereby a outputof the second filter is subtracted from an output of the first filter; anon-linear function having an output terminal connected to an inputterminal of the second filter; the non-linear function and the secondfilter being connected to receive a same RF signal as inputs.
 22. Thetransmitter of claim 21, further comprising a phase shifter having anoutput terminal connected to an input terminal of the auxiliary poweramplifier; a second non-linear function having an output terminalconnected to an input terminal of the phase shifter; wherein the same RFsignal is applied to each of the first filter, the first non-linearfunction, and the second non-linear function.
 23. The transmitter ofclaim 7, wherein the means for emulating and compensating for thenon-linear behavior of the output current of said auxiliary poweramplifier in the input signal to said main amplifier comprises a firstnon-linear function having an output terminal connected to an inputterminal of the main amplifier, and further comprising: a distortioncanceling filter having an output terminal connected to an inputterminal of the auxiliary power amplifier; a second non-linear functionhaving an output terminal connected to an input terminal of thedistortion canceling filter; the first non-linear function and thesecond non-linear function being connected to receive a same RF signalas inputs.
 24. The transmitter of claim 7, wherein the means foremulating and compensating for the non-linear behavior of the outputcurrent of said auxiliary power amplifier in the input signal to saidmain amplifier comprises: a first filter having an output terminalconnected to an input terminal of the main amplifier; a first non-linearfunction having an output terminal connected to an input terminal of thefirst filter; a second filter having an output terminal connected to aninput terminal of the auxiliary power amplifier; a second non-linearfunction having an output terminal, connected to an input terminal ofthe second filter; the first non-linear function and the secondnon-linear function being connected to receive a same RF signal asinputs.